整流器论文中英文资料外文翻译文献

1、整流器论文中英文资料外文翻译文献AC Voltage and CurrentSensorlessControl ofThree -Phase PWM Rectifiers1 THREE-PHASE PWM RECTIFIERSA SystemModelingFig. 1 shows the power circuit of the three-phase PWM rectifier. The voltage equationsaregiven byeaR pL00i avaeb0R pL0i bvbec00R pL i cvc( 1)Fig. 1. Three-phase PWM rectif

2、ier without ac-side sensors.where , and are the sourcevoltage, the line current, andthe rectifier input voltage, respectively and are the input resistance and the input inductance, respectively. When the peakline voltage , angular frequency , and initial phase angle are given, assuming a balanced th

3、ree-phase system, the source phase voltage is expressedaseacos2ebE cos()3eccos(2)3( 2)Where1t0(3)A transformation matrix based on the estimated phase angle ,which transformsthree-phasevariablesinto a synchronous dq referenceframe, iscos2C3sinMcos( M32)cos( M23)Msin( M32)sin( M32)( 4)Transforming ( 1

4、) into the referenceframe using ( 4)eqcR pLM LiqcvqcedcM LR pLidcvdc( 5)where p is adifferential operatorand . MMExpressing( 5) in a vector notationeRiM LJipLi v( 6)where,eqciqcvqc0 1eivJedc ,idc ,vdc ,1 0( 7)Taking a transformation of ( 2) by using ( 4)E coseE sin( 8)WhereM( 9)Expressing ( 6) and (

5、 8)ina discrete domain, by approximating thederivative term in ( 6) by a forward difference 9, respectively,e(k 1)Ri( k1)M LJi (k 1)Li (k1)v( k1)i (k )( 10)TE cos(k1)e( k 1)E sin(k1)( 11)2Where T is the sampling period.Fig. 2. Overall control block diagram.B SystemControlThe PI controllers are used

6、to regulate the dc output voltage and the acinput current. For decoupling current control, the cross-coupling terms arecompensatedin a feed forward-typeand the source voltage is also compensatedas a disturbance. For transient responseswithout overshoot, the anti-windup technique is employed 10. Theo

7、verall control block diagram eliminating the sourcevoltage and line current sensorsis shown in Fig. 2. The estimationalgorithm of sourcevoltages andline currentsis describedin the following sections.2 PREDICTIVE CURRENT ESTIMATIONThe currentsof I a ( k) and I c (k ) can not be calculated instantly s

8、ince thecalculation time of the DSP is required To eliminate the delay effect, a stateobserver can be used In.addition, the state observer provides the filtering.effectsfor the estimatedvariable.Expressing5 in a state spaceform,( )-g12x AxBu()yCx(13)where,3R1L0ABL10R1CL ,0,01Lxiqcueqcvqcidcedcvdc,An

9、d y is the output.Transforming ( 12) and( 13)into a discretedomain, respectively,X (k1)FX ( k)GU ( k)( 14)Y (k)HX ( k)(15)where,1RT1 TT0FLGLRT1TT01L,LThen, the observerequationadding anerror correction term to is given byGU (k )K (Y (k)( 16)X (k 1)F X (k)Y (k )Where K is the observer gain matrixand

10、“ ” means the estimated1) is the statevariable estimatedaheadone sampling period.quantity,and X (kSubtracting ( 15)from( 16) , the error dynamic equation of the observer isexpressedaserr (k1) FKC err (k )(17)whereerr (k ) X (k). Here, it is assumedthat the model parametersmatchX (k )well with the re

11、al ones. Fig. 3 showsthe block diagram of the closed-loop state observer.The state variable error depends only on the initial error and is independentof the input . For ( 17) to convergeto the zero state,the rootsof the characteristicequationof ( 17) should belocatedwithin the unit circle.Fig. 3. Cl

12、osed-loop state observer.Fig. 4. Short pulse region.4 EXPERIMENTS AND DISCUSSIONSA. SystemHardware ConfigurationFig. 5 showsthe systemhardwareconfiguration. The sourcevoltage isa three-phase, 110 V. The input resistanceand inductance are 0.06and 3.3 mH, respectively. The dc link capacitance is 2350F

13、and the switching frequency of the PWM rectifier is 3.5 kHz.5Fig. 5. System hardware configuration.Fig. 6. Dc link currents and corresponding phase currents ( in sector V ) .The TMS320C31 DSP chip operating at 33.3 MHz is used as a main processorand two 12-b A/D convertersare used. One of them is de

14、dicatedfor detecting the dc link current and the other is usedfor measuringthe dc output6voltage and the sourcevoltages and currents, where ac side quantities are just measuredfor performancecomparison.One of two internal timers in the DSP is employed to decide the PWM control period and the other i

15、s usedto determinethe dc link current interrupt. Considering the rectifier blanking time of 3.5 s, A/D conversion time of 2.6 s, andthe other signal delay time, the minimum pulse width is setto 10 s.A. Experimental ResultsFig. 6 showsmeasureddc link currentsand phasecurrents. In caseof sectorV of th

16、e space vector diagram, the dc link current corresponds to for the switching state of and for that of . Fig. 7( a) shows the raw dc link current before filtering . It has a lot of ringing componentsdue to the resonanceof the leakageinductanceandthe snubbercapacitor. When the dc current is sampled at

17、 the end point of the active voltage vectors as shown in the figure, the measuringerror canbereduced.Fig. 7. Sampling of dc link currents.7Fig. 8. Estimated source voltage and current at starting.To reduce this error further, the low passfilter should be employed, of which result is shown in Fig. 7(

18、 b) . The cut-off frequency of the Butterworths second-order filter is 112 kHz and its delay time is about 2 sec. Since the ringing frequency is 258 kHz and the switching frequency is 3.5 kHz, the filtered signal without significant delay is acquired.Fig. 8 shows the estimatedsourcevoltage and curre

19、nt at starting. With the proposedinitial estimation strategy, the starting operation is well performed. Fig. 9 showsthe phaseangle,magnitude, andwaveform of the estimatedsource voltage, which coincide well with measuredones.Fig. 10 showsthe source voltage and current waveform at unity power factor.

20、Figs. With the estimated quantities for the feedback control, the control performanceis satisfactory. The dc voltage variation for load changeswill be remarkably decreasedif a feedforward control for theload current is added,which is possible without additional cur-rent sensor when the PWM rectifier

21、 iscombined with the PWM inverter for ac motor drives.8Fig. 9. Estimated source voltage in steady state.( a) phase angle ( b) magnitude ( c) waveform.Fig. 10. Source voltage and current waveforms.(a) estimated ( b) measured.4 CONCLUSIONSThis paper proposed a novel control scheme of the PWM rectifier

22、s without employing any ac input voltage and current sensorsand with using dc voltage and current sensorsonly. Reducing the number of the sensorsused decreasesthe systemcost aswell asimproves the systemreliability . The phase angle and the magnitude of the source voltage have been estimated by contr

23、olling the deviation betweenthe rectifier current and its model current to9be zero. For line current reconstruction,switching statesand measureddc link currents were used. To eliminate the effect of the calculation time delay of the microprocessor,the predictive stateobserver was used. It was shown

24、that the estimation algorithm is robust to the parametervariation. The whole algorithm has been implemented for a proto-type 1.5 kVAPWM rectifier system controlled by TMS320C31 DSP. The experimental results have verified that the proposedac sensorelimination method is feasible.无交流电动势、电流传感器的三相PWM 整流器

25、控制1 三相 PWM 整流器10A 系统模型图一所示为三相 PWM 整流器的主电路，电压等式给出如下：eaR pL00i avaeb0R pL0i bvbec00R pL i cvc( 1)图 1无交流传感器三相PWM 整流器其中 e，i 和 v 分别是源电压，线电流和整流器的输入电压，R 和 L 分别是输入电阻和输入电感。当已知线电压峰值 E，角频率 和初始相位角 时，假定三相系统是平衡的，则源相位电压可以表达为eacos2ebEcos()3eccos(2)3( 2)其中t0( 3)一种基于估计相位角m 的变换矩阵，将三相变量变换成一个同步的，d q 坐标系，这个矩阵是cos Mcos( M

26、32)cos( M2)23C3sin(32)sin( M32)sinMM(4)将（ 1）式变为 dq 坐标系使用式（ 4）eqcRpLM LiqcvqcedcM LRpLidcvdc(5)其中 p 是一个微分算子且MM11将（ 5）式写成矢量形式eRiM LJipLiv其中eqciqcvqc01eivJedc ,idc ,vdc ,10用式（ 4）对（ 2）式进行变换E coseE sin其中M通过前向差分来接近微分的限幅，分别将（6）式和（ 8）式用离散域表示e(k 1)Ri( k1)M LJi (k 1)Li (k1)v(k1)i (k )TE cos(k1)e(k 1)E sin(k1)

27、其中， T 是采样周期( 6)( 7)( 8)( 9)( 10)( 11)图 2 总的控制模块图B 系统控制PI 控制器是用来调节直流输出电压和交流输入电流的。对于解耦电流控制，交叉耦合项用前馈式补偿，同时，源电压作为扰动的补偿。对于没有过调的暂态响应，引入 anti-windup 技术。消除源电压和线电流传感器的总的控制模块图如图2 所示。源电压和线电流的估计算法在以后的章节中介绍。122 预测电流估计由于 DSP 存在计算时间，所以 I a (k ) 和 I c ( k) 不能立即计算。为了消除延迟的影响，可以使用状态监测器。另外，状态监测器可以对估计变量起到滤波作用。将式（ 5）用状态空

28、间形式表达为gx AxBu(12)yCx(13)其中R1L0ABL10R1L ,0,CL01iqceqcvqcxuidc,edcvdcY 是输出。分别将式（ 12）和式（ 13）分别变换成离散领域X ( k1)FX (k )GU (k )( 14)Y(k) HX (k )( 15)其中1RT1TT0FLRGLT1T10T,LL则加入了误差调整的监测器等式为GU (k )K (Y(k )( )X ( k1) F X (k)Y( k)16其中， k 是监测器增益矩阵，“是提前一个采样周期”是指估计量， X (k 1)估计的状态变量。用式（15）和减去式（ 16），监测器的动态误差等式表述为err

29、(k 1) FKC err (k )( 17)其中 err (k ) X (k)7 所示X (k) 这里，假设模型参数与真实系统吻合的很好。图是闭环状态监测器的模块图。13状态变量误差仅取决于初始误差，与输入无关。为了使式（ 17）趋于零状态，典型等式（ 17）的根应该限制在单位圆内。图 3闭环状态监测器图 4 短脉冲区域3 实验与讨论A 系统硬件构造14图 5系统硬件结构图 6 直流电流和相应相电流(扇区 5).图 5所示是系统的硬件结构图。源电压是三相110V。输入电阻和电感分别为0.06和 3.3mH。直流侧电容为 2350F，PWM 整流器的开关切换频率为 3.5KHZ . 使用 TMS320C31